Signal processing method and apparatus for a spread spectrum radio communication receiver

ABSTRACT

A signal received by a base station is fed to a set of filters matched to spreading codes allocated to pilot channels originating from radio terminals. The outputs from the matched filters are processed so as to estimate parameters comprising, for each channel, eigenvectors representing propagation paths associated with respective amplitudes of reception along these paths. The processing of the output signals from the matched filters comprises an estimation of parameters for a first channel received with a fairly high power by the base station, a correction of the output from the matched filter of a second channel having less energy, taking account of at least one of the eigenvectors of the first channel, and an estimation of parameters for this second channel on the basis of the modified output from said matched filter.

BACKGROUND OF THE INVENTION

[0001] The present invention relates to mobile radio communicationsystems using code division multiple access (CDMA) techniques. It lieswithin the receivers used in the fixed stations of these systems andoperating a coherent demodulation of spread spectrum signals originatingfrom a set of radio terminals.

[0002] Coherent demodulation requires various parameters representingthe propagation channel between the transmitter and the receiver. Someof these parameters vary relatively slowly and can be estimated bystatistical survey procedures. Such is the case for example for thedelays assigned to the multiple propagation paths in the conventionalrake receiver. The delays specific to the various paths can be updatedat fairly low frequency, for example of the order of about a hundredmilliseconds. Other parameters have abrupt variations, on the scale ofthe duration of an information symbol, which are due to the phenomenonof fading. Such is the case in particular for the instantaneousamplitudes of reception of the symbols following the propagation pathstaken into consideration, which are necessary for coherent demodulation.These instantaneous amplitudes are complex amplitudes, manifesting theattenuation and the phase shift which are undergone at each instantalong the paths. In general, these complex amplitudes are estimated fromsymbols known a priori, or pilot symbols, interspersed among theinformation symbols transmitted so as to allow coherent demodulation.

[0003] It is known that CDMA receivers are affected by the near-farproblem according to which transmission from mobiles near a base stationtends to mask that from distant mobiles. The reason is that all thesemobiles share the same uplink frequency at the same time, thedistinction between them resulting from the quasi-orthogonality of thespreading codes. Transmission power controls procedures are implementedin order to limit the impact of this problem, but nonetheless the basestation does not receive with the same power the signals transmitted bythe various mobiles, in particular when the spreading factors of thevarious channels are not the same or when a mobile is moving fast, whichmay bias the power control algorithm.

[0004] In general, channel estimation methods make the assumption thatthe noise present on a channel is white noise. This assumption iscorrect in the case of thermal noise, although not for the noisegenerated by the other transmitters. The estimation methods thengenerate errors which are especially appreciable when the power receivedon a channel dominates that received on one or more other channels.

[0005] An object of the present invention is to make the estimates ofsuperposed CDMA channels more reliable in order to improve theperformance of the receivers.

SUMMARY OF THE INVENTION

[0006] The invention thus proposes a signal processing method for a basestation of a code division multiple access radio communication system,wherein a first signal received by the base station is fed to a set offilters matched to spreading codes allotted to pilot channelsoriginating from respective radio terminals, and output vectors from thematched filters are processed so as to estimate parameters representingsaid channels. The estimated parameters for a channel originating from aradio terminal comprise eigenvectors representing propagation pathsbetween this radio terminal and the base station, respectivelyassociated with amplitudes of reception along said paths. According tothe invention, the processing of the output vectors from the matchedfilters comprises an estimation of parameters for at least one firstchannel on the basis of the output vector from a first filter matched tothe spreading code allocated to said first channel, a modification ofthe output vector from a second filter matched to the spreading codeallocated to at least one second channel, the output vector from thesecond matched filter having on average less energy than the outputvector from the first matched filter, and an estimation of parametersfor said second channel on the basis of the modified output vector ofthe second matched filter. The modification of the output vector fromthe second matched filter comprises at least one subtraction of a vectorproportional to a projection of said output vector from the secondmatched filter parallel to a vector of the form M^(H).M₁.v_(1,i), whereM₁ and M are matrices determined by the spreading codes respectivelyallocated to said first and second channels, v_(1,i) is one of theestimated eigenvectors for the first channel, and (.)^(H) designates theconjugate transpose.

[0007] The eigenvector v_(1,i) is advantageously the estimatedeigenvector for the first channel which is associated with the amplitudewhose modulus is on average the highest. It is also possible to takeaccount of several eigenvectors associated with eigenvalues of highmodulus in the diagonalization of the autocorrelation matrices of thefading on the various channels.

[0008] A correction of the estimated impulse responses of channelsreceived relatively weakly is thus undertaken, after the matchedfiltering, so as to take account of the interference caused by one ormore paths of one or more channels received with more power.

[0009] This correction does not introduce very much complexity into thereceiver, especially when short spreading codes are used.

[0010] It amounts to projecting the estimated impulse response vector ofthe low-energy channel (at the output of the matched filter) onto asubspace orthogonal to the eigenvector associated with the strongestpath of the channel received with the most power. The bias thusintroduced is very weak and it avoids the estimation errors due to thepossible presence of a powerful interferer.

[0011] To avoid any problems of noise amplification when the white noiseis more powerful than the interferer, a weighting of the projection ispreferably undertaken. The subtracted vector is then given by${\frac{\alpha_{i}}{\alpha_{i} + {N0}}{J_{i} \cdot J_{i}^{H} \cdot \hat{h}}},$

[0012] where ĥ is the output vector from the second matched filter,${J_{i} = {\frac{1}{Q} \cdot M^{H} \cdot M_{1} \cdot v_{1,i}}},$

[0013] Q is the spreading factor on the pilot channels, α_(i) is anaverage power of reception along the propagation path represented by theeigenvector v_(1,i) (eigenvalue associated with v_(1,i)) and N0 is anestimated noise power on the second channel.

[0014] The method is applicable when the signals are transmitted on twoparallel pathways or channels between the transmitter and the receiver,for example two quadrature pathways, one of which comprises pilotsymbols known a priori and the other of which comprises unknowninformation symbols. This case is that of the uplink in third-generationcellular systems of UMTS (“Universal Mobile Telecommunications System”)type.

[0015] Each of the pilot channels is then formed jointly with aquadrature data channel emanating from the same radio terminal. A secondsignal, received by the base station in phase quadrature with respect tosaid first signal, is fed to a second set of filters matched tospreading codes respectively allocated to the data channels originatingfrom the radio terminals. Some at least of the output vectors from thematched filters of the second set are modified, and information symbolstransmitted on a data channel originating from a radio terminal areestimated by feeding the modified output vector from the second filtermatched to the spreading code allocated to this data channel to a rakereceiver defined by the parameters estimated for the pilot channeloriginating from said radio terminal. The modification of the outputvector from said matched filter of the second set comprises at least onesubtraction of a vector proportional to a projection of said outputvector parallel to a vector of the form M′^(H).M₁.v_(1,i), where M′ is amatrix determined by the spreading code allocated to said data channel.

[0016] Stated otherwise, the same correction by projection is performedon the pilot channels and on the data channels.

[0017] Another aspect of the present invention relates to a signalprocessing device for a base station, tailored to the implementation ofthe above method.

BRIEF DESCRIPTION OF THE DRAWINGS

[0018]FIG. 1 is a block diagram of the reception part of a base stationaccording to the invention.

[0019]FIGS. 2 and 3 are schematic diagrams of receivers belonging to thebase station of FIG. 1.

DESCRIPTION OF PREFERRED EMBODIMENTS

[0020] The invention is described below within the framework of a spreadspectrum radio communication system using a code-division multipleaccess technique (CDMA) of which UMTS is an example. A channel of such asystem on a carrier frequency is defined by a spreading code composed ofdiscrete samples called “chips”, having real values (±1) or complexvalues (±1±j)/{square root}{square root over (2)}, which follow oneanother at a chip rate F_(c).

[0021] In the case of the uplink (from the terminals to the basestations) of a UMTS network in frequency division duplex (FDD) mode, aterminal uses two quadrature channels each using a real-valued spreadingcode together with binary phase modulation (BPSK, “Binary Phase ShiftKeying”) and F_(c)=3.84 Mchip/s. The two BPSK signals modulate twoquadrature radio waves. One of these two channels (I pathway) carriesthe user data, and the other (Q pathway) carries control information.

[0022] This control information comprises pilot bits known a priori tothe base station and which allow it to estimate the parameters of thepropagation channel. In what follows, reference will therefore be madeto the I pathway as being the pilot channel.

[0023]FIG. 1 shows the reception part of a base station (“node B”) of aUMTS type network in FDD mode operating on the two quadrature pathways(I and Q). The I pathway (real part of the complex baseband signal)transports the data bits, while the Q pathway (imaginary part)transports the control bits, in particular the pilot bits, with forexample a spreading factor Q=256. For a precise description of theseuplink channels reference may be made to the technical specification 3GTS 25.211, version 3.3.0, “Physical Channels and Mapping of TransportChannels onto Physical Channels (FDD) (Release 1999)”, published in June2000 by the 3GPP (“3^(rd) Generation Partnership Project”), section5.2.1.

[0024] The base station illustrated by FIG. 1 comprises a radio stage 1which performs the analogue processing required on the radio signalpicked up by the antenna 2. The radio stage 1 delivers a complexanalogue signal whose real and imaginary parts are digitized by theanalogue/digital converters 3 on respective processing pathways I and Q.On each pathway, a matched filter 4 tailored to the pulse shaping by thetransmitter produces a digital signal r_(I), r_(Q) at the chip rate ofthe spreading codes.

[0025] The resultant complex signal r=r_(I)+j.r_(Q) is fed to receivers5 which effect the processing for estimating the responses of k channelsand the symbols sent on these k channels.

[0026] The signals transmitted by a terminal to a base station propagatealong multiple paths, and arrive at the base station together withsignals sent by other terminals with other spreading codes.

[0027] Consider the reception by the base station of an unknowninformation bit b on the I pathway from a mobile terminal, insynchronism with a known pilot bit (equal to 1 for example) on the Qpathway. The duration 1/F_(S) of a symbol (bit) on the channel is amultiple of the chip duration, the ratio of the two being the spreadingfactor Q=F_(C)/F_(S) of the channel. In the example of UMTS, thespreading factor Q is a power of 2 lying between 4 and 256.

[0028] Moreover, L denotes the number of propagation paths allowed forby the receiver 5, and W the length of the impulse response of thechannel, expressed in terms of number of chips (for example W=6.Q for aresponse over 400 μs).

[0029] The receiver 5 uses a “rake” receiver 10 of conventional type(FIGS. 2 and 3). The channel propagation profile is defined by a set ofeigenvectors v_(i) and of associated eigenvalues λ_(i) for 0≦i<L, whichis calculated with a relatively large periodicity (for example of theorder of about a hundred milliseconds) by a channel analysis module 11.Each eigenvector v_(i), of dimension W, is a waveform associated with anecho in the impulse response of the channel. In a traditional “rake”receiver, each eigenvector v_(i) represents a pulse suffering a timeshift. In the “rake” receiver 10, each eigenvector v_(i) is associatedwith an amplitude a_(i) which varies from one bit to another. Theamplitude a_(i), calculated for each bit by the module 11, is a randomvariable such that the mathematical expectation of |a_(i)|² is equal tothe eigenvalue λ_(i). The “rake” receiver 10 receives an input vector ofdimension W, calculates the scalar product of this input vector witheach of the eigenvectors v_(i), then the sum of these L scalar productsweighted by the amplitudes a_(i)*. This weighted sum is a soft estimateof the bit sent b.

[0030] In a manner known per se, the channel analysis module 11 canproceed as follows in order to estimate the pairs of parameters (v_(i),a_(i)) from successive input vectors X of dimension W representingsuccessive estimates of the impulse response of the relevant channelwhich are obtained by means of the pilot bits:

[0031] calculation of the mathematical expectation K of the matrixX.X^(H) over a typical duration of the order of about a hundredmilliseconds;

[0032] diagonalization of the matrix K, and selection of the Leigenvalues of largest moduli λ_(i) (0≦i<L, with |λ₀|≧|λ₁|≧ . . .≧|λ_(L−1)|) respectively associated with eigenvectors v_(i);

[0033] projection of the vector X onto each of the eigenvectors retainedv_(i), so as to obtain the amplitudes a_(i). These amplitudes may beobtained by simple projection (a_(i)=v_(i) ^(H).X), or by weightedprojection so as to produce the estimate according to the maximum aposteriori criterion$\left( {{{MAP}\text{:}\quad a_{i}} = {\frac{\lambda_{i}}{\lambda_{i} + {N0}} \cdot v_{i}^{H} \cdot X}} \right),$

[0034] N0 designating an estimate of the noise power on the channel).Other known estimation procedures are also useable (estimation in thesense of least squares, under zero constraint, etc.);

[0035] calculation of the noise power N0 equal to the mathematicalexpectation of the energy per bit |B|² of the residual noise$B = {X - {\sum\limits_{i = 0}^{L - 1}{a_{i} \cdot {v_{i}.}}}}$

[0036] In regard to the terminal handled by a receiver 5, the signal rreceived by the base station in respect of the relevant bit b can bewritten in the form of a vector of dimension Q+W−1:

r=(b.M′+j.M).h+n  (1)

[0037] where:

[0038] ĥ is a vector of dimension W containing the impulse response ofthe channel between the terminal and the base station, sampled at thechip rate;

[0039] M and M′ are Toeplitz matrices with Q+W−1 rows and W columns,whose columns are defined by the spreading codes c=[c(0), c(1), . . . ,c(Q−1)] and c′=[c′(0), c′(1), . . . , c′(Q−1)] used by the terminal onthe Q pathway and on the I pathway, respectively: $\begin{matrix}{M = \begin{pmatrix}{c(0)} & 0 & \ldots & 0 \\{c(1)} & {c(0)} & ⋰ & \vdots \\\vdots & {c(1)} & ⋰ & 0 \\{c\left( {Q - 1} \right)} & \vdots & ⋰ & {c(0)} \\0 & {c\left( {Q - 1} \right)} & \quad & {c(1)} \\\vdots & ⋰ & ⋰ & \vdots \\0 & \ldots & 0 & {c\left( {Q - 1} \right)}\end{pmatrix}} & (2) \\{M^{\prime} = \begin{pmatrix}{c^{\prime}(0)} & 0 & \ldots & 0 \\{c^{\prime}(1)} & {c^{\prime}(0)} & ⋰ & \vdots \\\vdots & {c^{\prime}(1)} & ⋰ & 0 \\{c^{\prime}\left( {Q - 1} \right)} & \vdots & ⋰ & {c^{\prime}(0)} \\0 & {c^{\prime}\left( {Q - 1} \right)} & \quad & {c^{\prime}(1)} \\\vdots & ⋰ & ⋰ & \vdots \\0 & \ldots & 0 & {c^{\prime}\left( {Q - 1} \right)}\end{pmatrix}} & (3)\end{matrix}$

[0040] n is a vector containing samples of additive noise. This noise isnot necessarily white noise since it incorporates the contributions fromthe other channels, emanating from the other terminals transmitting atthe same time.

[0041] On the I pathway, the real part r_(I) of this signal r is fed toa matched filter 12 corresponding to the spreading code c′ assigned tothe data bits of the channel. On the Q pathway, another matched filter13, operating with the spreading code c, receives the imaginary partr_(Q) of the signal r and produces a first estimate of the impulseresponse of the channel:

ĥ=M ^(H) .r _(I)  (4)

[0042]FIG. 2 uses similar notation to that of FIG. 3. An index 1 isappended thereto to signify that the receiver of FIG. 2 is the one whichprocesses the channel having the most energy. This channel is easilyidentified, for example by maximizing the average of the norm of thevectors ĥ over the period of updating of the eigenvectors by the modules11.

[0043] In this receiver 5 assigned to the most powerful channel (FIG.2), the channel analysis module 11 operates on the basis of the vectorsX=ĥ₁ successively supplied by the matched filter 13. It deducestherefrom the eigenvectors v_(1,i) and the associated instantaneousamplitudes a_(1,i) (0≦i<L) supplied to the “rake” receiver 10. Theeigenvalues corresponding to the eigenvectors v_(1,i) are denotedα_(i)=λ_(1,i).

[0044] The highest-energy path of the response of this most powerfulchannel corresponds to the eigenvector v_(1,0) and to the eigenvalueλ_(1,0)=α₀. This eigenvector v_(1,0) defines in the signal space adirection along which interference may affect the other channels. Afterprojection onto the signal subspace corresponding to another channelafter matched filtering, the direction of interference is defined by thenormed vector: $\begin{matrix}{{J_{0} = {\frac{1}{Q} \cdot M^{H} \cdot M_{1} \cdot v_{1,0}}}{{{for}\quad a\quad Q\quad {pathway}},}} & (5) \\{J_{0}^{\prime} = {\frac{1}{Q} \cdot M^{\prime \quad H} \cdot M_{1} \cdot v_{1,0}}} & (6)\end{matrix}$

[0045] for an I pathway, where M₁ is the matrix of codes which relatesto the pilot channel received with the most power, taking account of thepossible time shift δ (in terms of number of chips) between the relevantchannel and the most powerful channel: $\begin{matrix}{M_{1} = \begin{pmatrix}{c_{1}(\delta)} & 0 & \ldots & 0 \\{c_{1}\left( {\delta + 1} \right)} & {c_{1}(\delta)} & ⋰ & \vdots \\\vdots & {c_{1}\left( {\delta + 1} \right)} & ⋰ & 0 \\{c_{1}\left( {\delta + Q - 1} \right)} & \vdots & ⋰ & {c_{1}(\delta)} \\0 & {c_{1}\left( {\delta + Q - 1} \right)} & \quad & {c_{1}\left( {\delta + 1} \right)} \\\vdots & ⋰ & ⋰ & \vdots \\0 & \ldots & 0 & {c_{1}\left( {\delta + Q - 1} \right)}\end{pmatrix}} & (7)\end{matrix}$

[0046] The matrices${\frac{1}{Q} \cdot M^{H} \cdot M_{1}}\quad {and}\quad {\frac{1}{Q} \cdot M^{\prime \quad H} \cdot M_{1}}$

[0047] involved in the expressions (5) and (6) are constant when shortcodes are used (periodicity of one bit time), thereby minimizing thecalculations. Otherwise, the vectors J₀ and J′₀ need to be calculatedfor each bit.

[0048] These two matrices are calculated by respective modules 15 and 16in the receivers of the type represented in FIG. 3, as a function of thecodes c and c₁ and the shift δ for $\frac{1}{Q} \cdot M^{H} \cdot M_{1}$

[0049] (module 15) and as a function of the codes c′ and c₁ and of theshift δ for $\frac{1}{Q} \cdot M^{\prime \quad H} \cdot M_{1}$

[0050] (module 16). In each receiver (FIG. 3), the eigenvector v_(1,0)is multiplied by the matrices $\frac{1}{Q} \cdot M^{H} \cdot M_{1}$

[0051] and $\frac{1}{Q} \cdot M^{\prime \quad H} \cdot M_{1}$

[0052] produced by the modules 15 and 16 so as to obtain the normedvectors J₀ and J′₀ according to relations (5) and (6), respectively.

[0053] A module 16 is also present in the receiver of FIG. 2 so as tocalculate the matrix$\frac{1}{Q} \cdot M_{1}^{\prime \quad H} \cdot M_{1}$

[0054] as a function of the codes c′₁ and c₁ (the shift δ is zerobetween the I and Q pathways which are synchronous). This matrix servesalso to determine an interferer direction J′₀ for the data pathway.

[0055] In the receiver according to FIG. 3, the interference directionJ₀ is supplied to a module 17 which projects the estimate ĥ of theimpulse response delivered by the matched filter 13 onto a subspaceorthogonal to J₀, thereby giving rise to a corrected response vectorH^(⊥). The correction consists more precisely in deducting from ĥ avector proportional to its projection along the direction J₀. We can inparticular take: $\begin{matrix}{h^{\bot} = {{\hat{h} - {\frac{\alpha_{0}}{\alpha_{0} + {N0}}{J_{0} \cdot J_{0}^{H} \cdot \hat{h}}}} = {\left\lbrack {{Id} - {\frac{\alpha_{0}}{\alpha_{0} + {N0}}{J_{0} \cdot J_{0}^{H}}}} \right\rbrack \cdot \hat{h}}}} & (8)\end{matrix}$

[0056] where the weighted projection matrix$\left\lbrack {{Id} - {\frac{\alpha_{0}}{\alpha_{0} + {N0}}{J_{0} \cdot J_{0}^{H}}}} \right\rbrack$

[0057] is updated at relatively low frequency. It is the correctedvector h^(⊥)=X which is supplied to the channel analysis module 11.

[0058] Likewise, in each receiver, the interference direction J′₀ issupplied to a module 18 which projects the output vector y from thematched filter 12 onto a subspace orthogonal to J′₀, thereby giving riseto a corrected vector y′. The correction consists in deducting from y avector proportional to its projection along the direction J′₀. We can inparticular take: $\begin{matrix}{y^{\bot} = {{\hat{y} - {\frac{\alpha_{0}}{\alpha_{0} + {N0}}{J_{0}^{\prime} \cdot J_{0}^{\prime \quad H} \cdot \hat{y}}}} = {\left\lbrack {{Id} - {\frac{\alpha_{0}}{\alpha_{0} + {N0}}{J_{0}^{\prime} \cdot J_{0}^{\prime \quad H}}}} \right\rbrack \cdot \hat{y}}}} & (9)\end{matrix}$

[0059] where the weighted projection matrix$\left\lbrack {{Id} - {\frac{\alpha_{0}}{\alpha_{0} + {N0}}{J_{0}^{\prime} \cdot J_{0}^{\prime \quad H}}}} \right\rbrack$

[0060] is updated at relatively low frequency. It is the correctedvector y^(⊥) which is supplied to the “rake” receiver 10 to estimate theinformation symbol sent.

[0061] In the foregoing description, the projection of the vectors ĥ andŷ is performed onto a subspace orthogonal to a single interferencedirection J₀ or J′₀. It will be noted that it is possible to extend theprocedure to several interference directions defined by severalenergy-containing paths (within the limit of the dimension of thesubspace).

[0062] For example, for p>1 paths, we can take the p eigenvectorsv_(1,0), v_(1,1), . . . , v_(1,p−1) corresponding to the eigenvalues oflargest moduli α₀=λ_(1,0), α₁=λ_(1,1), . . . , α_(p−1)=λ_(1,p−1)identified by the analysis module 11 processing the response ĥ₁, and wecan determine the normed vectors: $\begin{matrix}{{{J_{i} = {\frac{1}{Q} \cdot M^{H} \cdot M_{1} \cdot v_{1,i}}}{for}\quad a\quad Q\quad {pathway}},\quad {and}} & (10) \\{J_{i}^{\prime} = {\frac{1}{Q} \cdot M^{\prime \quad H} \cdot M_{1} \cdot v_{1,i}}} & (11)\end{matrix}$

[0063] for an I pathway (0≦i<p). The weighted projection matrices ofrelations (8) and (9) are then replaced by$\left\lbrack {{Id} - {\sum\limits_{i = 0}^{p - 1}{\frac{\alpha_{i}}{\alpha_{i} + {N0}}{J_{i} \cdot J_{i}^{H}}}}} \right\rbrack$

[0064] and$\left\lbrack {{Id} - {\sum\limits_{i = 0}^{p - 1}{\frac{\alpha_{i}}{\alpha_{i} + {N0}}{J_{i}^{\prime} \cdot J_{i}^{\prime \quad H}}}}} \right\rbrack,$

[0065] respectively. The p relevant paths may also be identified byanalysis modules 11 belonging to distinct receivers.

[0066] The method is also applicable in the case where the base stationpossesses several distinct reception antennas, whose signals arecombined to afford space diversity. In a case with two antennas, thesubspace described by the main interferer (for a Q pathway) is generatedby the matrix: $\begin{matrix}{J_{s} = \begin{pmatrix}J_{0} & {{- \frac{\rho}{\rho }} \cdot J_{0}} \\{\frac{\rho}{\rho } \cdot J_{0}} & J_{0}\end{pmatrix}} & (12)\end{matrix}$

[0067] and the weighted projection matrix of relation (8) becomes:$\left\lbrack {{Id} - {J_{s} \cdot \begin{pmatrix}\frac{\alpha_{0} \cdot \left( {1 + {\rho }} \right)}{{\alpha_{0} \cdot \left( {1 + {\rho }} \right)} + {N0}} & 0 \\0 & \frac{\alpha_{0} \cdot \left( {1 - {\rho }} \right)}{{\alpha_{0} \cdot \left( {1 - {\rho }} \right)} + {N0}}\end{pmatrix} \cdot J_{s}^{H}}} \right\rbrack,$

[0068] where ρ is the correlation factor for the two antennas.

We claim:
 1. A signal processing method for a base station of acode-division multiple access radio communication system, comprising thesteps of: feeding a first signal received by the base station to a setof filters matched to spreading codes allotted to pilot channelsoriginating from respective radio terminals; and processing outputvectors from the matched filters to estimate parameters representingsaid channels, whereby the estimated parameters for a channeloriginating from a radio terminal comprise eigenvectors representingpropagation paths between said radio terminal and the base station,respectively associated with amplitudes of reception along said paths,wherein the processing of the output vectors from the matched filterscomprises: estimating parameters for at least one first channel on thebasis of the output vector from a first filter matched to the spreadingcode allocated to said first channel; modifying the output vector from asecond filter matched to the spreading code allocated to at least onesecond channel, the output vector from the second matched filter havingon average less energy than the output vector from the first matchedfilter; and estimating parameters for said second channel on the basisof the modified output vector of the second matched filter, and whereinthe modification of the output vector from the second matched filtercomprises at least one subtraction of a vector proportional to aprojection of said output vector from the second matched filter parallelto a vector of the form M^(H).M₁.v_(1,i), where M₁ and M are matricesdetermined by the spreading codes respectively allocated to said firstand second channels, v_(1,i) is one of the estimated eigenvectors forthe first channel, and (.)^(H) designates the conjugate transpose.
 2. Amethod according to claim 1, wherein said eigenvector v_(1,i) is theestimated eigenvector for the first channel which is associated with theamplitude having the highest modulus on average.
 3. A method accordingto claim 1, wherein the matrices M₁ and M are Toeplitz matrices havingcolumns given by the spreading codes respectively allocated to the firstand second channels, taking account of a time shift between the firstand second channels.
 4. A method according to claim 1, wherein saidsubtracted vector is given by${\frac{\alpha_{i}}{\alpha_{i} + {NO}}{J_{i} \cdot J_{i}^{H} \cdot \hat{h}}},$

where ĥ is the output vector from the second matched filter,${J_{i} = {\frac{1}{Q} \cdot M^{H} \cdot M_{1} \cdot v_{1,i}}},Q$

is a spreading factor on the pilot channels, α_(i) is an average powerof reception along the propagation path represented by the eigenvectorv_(1,i) and N0 is an estimated noise power on the second channel.
 5. Amethod according to claim 1, wherein each of said pilot channels isformed jointly with a quadrature data channel emanating from the sameradio terminal, the method further comprising the steps of feeding asecond signal, received by the base station in phase quadrature withrespect to said first signal, to a second set of filters matched tospreading codes respectively allocated to the data channels originatingfrom the radio terminals; modifying some at least of the output vectorsfrom the matched filters of the second set; and estimating informationsymbols transmitted on a data channel originating from a radio terminalby feeding the modified output vector from the second filter matched tothe spreading code allocated to said data channel to a rake receiverdefined by the parameters estimated for the pilot channel originatingfrom said radio terminal, and wherein the modification of the outputvector from said matched filter of the second set comprises at least onesubtraction of a vector proportional to a projection of said outputvector parallel to a vector of the form M′^(H).M₁.v_(1,i), where M′ is amatrix determined by the spreading code allocated to said data channel.6. A signal processing device for a base station of a code divisionmultiple access radio communication system, comprising: a set of filtersmatched to spreading codes allocated to pilot channels originating fromrespective radio terminals, to receive a first signal received by thebase station; and means for processing output vectors from the matchedfilters so as to estimate parameters representing said channels, wherebythe estimated parameters for a channel originating from a radio terminalcomprise eigenvectors representing propagation paths between said radioterminal and the base station, respectively associated with amplitudesof reception along said paths, wherein said processing means comprise:means for estimating parameters for at least one first channel on thebasis of the output vector from a first filter matched to the spreadingcode allocated to said first channel; means for modifying the outputvector from a second filter matched to the spreading code allocated toat least one second channel, the output vector from the second matchedfilter having on average less energy than the output vector from thefirst matched filter; and means for estimating parameters for saidsecond channel on the basis of the modified output vector of the secondmatched filter, and wherein the means for modifying the output vectorare arranged to subtract from the output vector from the second matchedfilter at least one vector proportional to a projection of said outputvector from the second matched filter parallel to a vector of the formM^(H).M₁.v_(1,i), where M₁ and M are matrices determined by thespreading codes respectively allocated to said first and secondchannels, v_(1,i) is one of the estimated eigenvectors for the firstchannel, and (.)^(H) designates the conjugate transpose.
 7. A deviceaccording to claim 6, wherein said eigenvector v_(1,i) is the estimatedeigenvector for the first channel which is associated with the amplitudehaving the highest modulus on average.
 8. A device according to claim 6,wherein the matrices M₁ and M are Toeplitz matrices having columns givenby the spreading codes respectively allocated to the first and secondchannels, taking account of a time shift between the first and secondchannels.
 9. A device according to claim 6, wherein said subtractedvector is given by${\frac{\alpha_{i}}{\alpha_{i} + {NO}}{J_{i} \cdot J_{i}^{H} \cdot \hat{h}}},$

where ĥ is the output vector from the second matched filter,${J_{i} = {\frac{1}{Q} \cdot M^{H} \cdot M_{1} \cdot v_{1,i}}},Q$

is a spreading factor on the pilot channels, α_(i) is an average powerof reception along the propagation path represented by the eigenvectorv_(1,i) and N0 is an estimated noise power on the second channel.
 10. Adevice according to claim 6, further comprising a second set of filtersmatched to spreading codes respectively allocated to data channelsformed jointly with the pilot channels originating from the radioterminals, so as to receive a second signal received by the base stationin phase quadrature with respect to said first signal, means formodifying some at least of the output vectors from the matched filtersof the second set, and means for estimating information symbolstransmitted on at least one data channel originating from a radioterminal, including a rake receiver defined by the parameters estimatedfor the pilot channel originating from said radio terminal and receivingthe modified output vector from the second filter matched to thespreading code allocated to said data channel, and wherein the means formodifying the output vector from a matched filter of the second setcomprise means for subtracting a vector proportional to a projection ofsaid output vector parallel to a vector of the form M′^(H).M₁.v_(1,i),where M′ is a matrix determined by the spreading code allocated to saiddata channel.